The present invention relates generally to electronic ballasts used to operate gas discharge lamps. More particularly, this invention pertains to circuits and methods used to control the preheating and ignition (“striking”) of a gas discharge lamp by an electronic ballast having a resonant tank output.
Conventional electronic ballasts typically combine a power factor correction (PFC) stage with a high frequency resonant inverter to preheat, strike and drive a fluorescent lamp at different frequencies. The parallel-loaded, series resonant inverter and LCC inverter (which has a smaller value of series-connected capacitors) are both widely used in electronic ballasts. FIG. 1 illustrates a simplified circuit for these inverter topologies driving a load of two series-connected lamps. Both circuit types have the same topology, but in the LCC version the blocking capacitor Cs is small enough that it contributes to the resonant properties instead of merely being a DC block. FIG. 1 also shows the filament preheat circuitry. The auxiliary windings L3, L4, and L5 are wound on the same core as inductor Lr to provide the preheat current to the lamp filaments. Capacitors C3, C4, and C5 present a lower impedance at the preheat frequency and a higher impedance at normal operating frequency to reduce filament loss after ignition of the lamp. Before lamp ignition, the resonant tank circuit comprising Lr and Cp dominates the behavior of the inverter, and a high voltage can be generated across Cp to strike the lamp. After lamp ignition, the impedance of the lamp is low such that Lr and Cs dominate the behavior of the circuit. The transfer functions of these circuits are well studied. Bode plots of the resonant tank circuit is plotted in FIG. 2, before and after the ignition of the lamp.
A conventional analog control circuit for an electronic ballast typically uses resistors to set three different inverter frequencies for preheating the filaments, striking the lamp, and operating the inverter at the normal running frequency. In such control circuits, the values of the resistors and capacitors can also be used to “program” the time duration of the preheat phase. These three inverter frequencies are plotted on FIG. 2 as points A, B, and C. Although there are limitations to programming these functions using different resistor and capacitor values, analog controllers are popular because of their low cost.
Other operational factors arise when the power flow of the inverter is considered. During normal ballast operation after ignition of the lamp, energy constantly circulates between Cp and Lr. As shown in FIG. 1, the current flowing in Lr (IL) is the sum of the lamp current (ILamp) and the current flowing through capacitor Cp (ICp). Because the voltage across the fluorescent lamps is determined by the lamp specification, ICp is a function of the value of Cp and the inverter frequency, which is generally between 40 kHz and 65 kHz. As an example, for an application having two T5 lamps connected in series, the AC voltage across Cp is approximately 250 V and the lamp current is 440 mA. The ratio of the currents ICp to ILamp is calculated over the range from 40 kHz and 65 kHz, with the value of Cp ranging from 1 nF to 4 nF. FIG. 3 shows that the ratio of the amplitudes of ICp to ILamp ranges from 0.4 to more than 0.9, with the Cp value between 3 nF and 4 nF. FIG. 3 also shows that ICp decreases significantly with smaller values of Cp and at lower frequencies. For a typical LCC tank, the currents IL, ICp and ILamp are illustrated as vectors in FIG. 4, where Vac is the vector of the fundamental frequency AC voltage of the output of the inverter and a is the angle between ILamp and ICp. The conduction loss of the current IL can be calculated with a geometric approach:R·IL2=R·ILamp2+R·ICp2+2R·|ILamp·ICp|cos(α)where R can be the resistance of either the inductor or the switches.
In a parallel loaded, series resonant inverter, because of the larger value of Cs, α is close to 90 degrees and the factor 2R·|ILampICp|cos(α) is very small. However, the R·ICp2 factor can still be high with a large value for Cp. For the LCC ballast circuit, ICp increases IL more significantly and with a being smaller, the conduction loss is even higher. In FIG. 4, the vectors of the voltages across the lamps, and across Cs, and Cp, are also shown at a different scale. Based on the phase relationship between the voltage and current of a capacitor,tan(α)=2τƒCs·Rlampwhere f is the normal running frequency and Rlamp is the resistance of the lamp, both the amplitude of ICp and α determine conduction loss. On the other hand, because the flux density of the core of the inductor is proportional to IL, a higher IL increases core losses in addition to the conduction loss.
In the lamp ignition phase, energy flows only into the resonant tank and builds up as current in Lr and voltage across Cp until the lamp starts to ignite. Thus, a high value Cp requires Lr to store more energy, which means either more losses or a larger core size. The peak voltage required to start the lamp is typically high and the components are subjected to the highest stress in this situation. With the load of the lamp removed from the circuit in FIG. 1, the inverter has only an LC tank as the load. Thus,
            1      2        ⁢          C      p        ⁢          V      AC_peak        ⁢    2    =            1      2        ⁢          L      r        ⁢                  ⁢          I      peak        ⁢    2  where the VAC—peak and Ipeak are the peak values of the AC voltage across Cp and the current in Lr.
With VAC—peak set by the lamp manufacturer to strike the lamp, and Lr set to provide a specified lamp current at the steady state frequency, Ipeak becomes a function of Cp:
      I    peak    =                              C          p                          L          r                      ⁢          V      AC_peak      
Obviously, Ipeak decreases with a reduced value of Cp. To avoid hard switching, Lr must not saturate at Ipeak. This requires a larger air gap with higher fringing losses, more winding turns with more conduction losses, and, in some cases, a bigger core with more core losses and higher cost.
Using a low value of Cp with traditional analog control circuits is not practical because of the stray capacitance associated with the connection between the ballast and the fixture and with the fixture itself. In the field, it is very common for the ballast output cable to connect to the lamps in the fixture after passing though 18 feet or more of conduit having a metal wrap. The stray capacitance from the ballast output cable to the conduit and to ground is effectively in parallel with Cp in the circuit, and is represented in FIG. 1 as Cstray. An example is shown in FIG. 5 for a LCC resonant tank with Lr=1.95 mH and Cs=15 nF. The value of Cp is selected to be low, 1.8 nF. Assuming Cstray varies from 0 to 200 pF, the frequency response of the striking voltage of the resonant tank before the ignition of lamp is illustrated in FIG. 5. With an increase in the stray capacitance or in the length of the external ballast output cable, the entire frequency response curve shifts to a lower frequency and the resonant frequency shifts from 85 kHz to 80.6 kHz. FIG. 6 shows the variation in measured peak lamp striking voltage as a function of the length of the conduit connected to the resonant tank, at a constant inverter frequency of 93 kHz. This measurement confirms that stray capacitance can result in insufficient striking voltage. Conventionally, analog ballasts for driving T8 and compact lamps are arranged to achieve ignition in the presence of a conduit by sweeping the ignition frequency. The frequency is steadily reduced, and eventually hits the resonant frequency and ignites the lamp. For linear lamp fixtures with the common connected filaments in parallel (the U.S. convention) the constraint on the use of this technique comes from the Underwriters Laboratory “through lamp leakage” requirement. This stipulates in effect a maximum duration for which a given ground fault current can persist. For T8 lamps this is on the order of 20 milliseconds, and it is just possible to execute a frequency sweep in this time. However, with T5HO lamps which run at much higher currents (440 mA instead of 180 ma) the permissible pulse duration is only about 1 millisecond and with current technology it is not possible to perform a frequency sweep during this time interval. Hence it becomes necessary to select the correct frequency for ignition for each length of conduit that is connected.
For most common filament heating circuitry as shown in FIG. 1, auxiliary windings are added to the same core of Lr, as L3 to L5 shown in FIG. 1, to provide the voltages to preheat the filaments. With external stray capacitance added to the tank, the frequency response curve shifts to the left, and the filament preheat voltage decreases. As the result, the filament preheat is not sufficient and the life span of the lamp is reduced. The conventional analog control chip used in electronic ballasts has very little flexibility and the only way to reduce the effects of stray capacitance is to increase the value of Cp.
Several approaches have been used in the prior art to address the problems of maintaining optimum lamp preheat and ignition conditions in microcontroller-based electronic ballasts. In one approach, a large resonant capacitor can be selected such that the affects of the stray capacitance associated with the output cable is small compared to the total resonant capacitance. In another approach, for instant start ballasts, during the start, the resonant inductor saturates. After saturation, the inductance value is very small. The resonant peak thus moves to a very high frequency, much higher than the striking frequency. Because the striking frequency is so far away from the resonant peak, the voltage on the resonant capacitor is no longer sensitive to the variation of the parameters of the resonant capacitor. This allows the ballast to start the lamp with different output cable lengths with essentially the same voltage. There are several obvious disadvantages to this solution. When such a ballast is in the lamp striking phase, it is operating deeply in a capacitive mode with high current and high voltage stresses on the inverter transistors. There can be more than 100 hard switching cycles when no lamp is connected, which is hazardous to the ballast.
In cases where the resonant inductor does not saturate, as seen in most program start ballasts, with a higher value of resonant capacitance and a lower lamp ignition voltage to start the lamp, it is not difficult to start the lamp. However, a higher resonant capacitance establishes a preheat frequency that cannot be much higher than the normal running frequency. As a result, the filament capacitor does not provide much attenuation to the filament current at normal operating frequency when under conditions when the preheat to the filaments is sufficient. The losses on the filaments are relatively high.
In either program start or instant start ballasts, a high value of the resonant capacitor results in high circulation current at steady state, which means higher conduction losses in the transistors and inductor.
What is needed, then, is an electronic ballast having a control circuit that can sense the operating environment of the ballast and adapt the ignition frequency of the inverter to provide optimum preheating and striking of the lamp connected to the ballast.